Radio Frequency (RF) Coupling networks are well known in the art and are used in many different applications. These applications include transmit power control, radar detection and control, isolation and feedback strategies. Particular to the portable radio, coupler designs are critical to optimizing transmit output power while providing protection to the transmit power amplifier. A properly designed coupler will provide differentiation between incident and reflected RF energy (directivity) while exhibiting uniform coupling efficiency over the desired RF frequency range. The physical design of the coupler must accommodate well understood relationships between RF voltage potentials and their associated electric fields, and RF current densities and its associated magnetic fields. Both magnetic (H-field) and electric fields (E-field) are present in a propagating RF signal; however, the coupler's physical design can be constructed to primarily operate off either field (electric or magnetic) as the fundamental coupling mechanism. The relationship between the two fields are defined by a family equations known as Maxwell's equations. The electromagnetic vector relationships germane to this discussion are shown in Table 1 below:
TABLE 1S = E × HwhereS =Power per unit area (W/m{circumflex over ( )}2)-Poynting VectorE =Electric Field Intensity VectorH =Magnetic Field Intensity VectorandV × H = J + ∂D/∂t;whereV =Direction of the axis of rotation and the magnitude of therotation VectorH =Magnetic Field Intensity VectorJ =Current Density VectorD =Magnetic Flux Density∂/∂t =time derivative
From Maxwell's equations, it is shown that a time varying electric and magnetic field generated by RF current densities or voltage potentials applied to a transmission line induce electromagnetic (EM) fields in the surrounding region. If a second conductor is positioned within the region, the magnetic field (H-field) will induce a current vector J proportional to the conductor's surface area which is perpendicular to propagating EM fields
The vector direction of the induced field is determined by the Poynting Vector S. This principle holds true for both co-planar and offset structures that are positioned on different planes from the main transmission line. By using the “Right Hand Rule” to S, the direction of the current flow within a given conductor can be determined by rotating the index finger of the right hand from the E-field vector to the H-field vector, and noting that the extended thumb points in the direction of the current flow. A diagram of a transmission line with a coupling structure positioned in an upper plane (not directly over) the main transmission line is shown in FIG. 1.
Prior art FIG. 1 shows the top surface of the transmission line 101 exhibits a magnetic field polarity that is “additive” to the H-field of the lower surface Upper Plate 100 (coupling structure). For this condition to exists, the induced current “io” in the upper plate 100 must flow in the OPPOSITE direction of the current “i1” in transmission line 101. If the plate current were “forced” to flow in the same direction as the transmission line current, the H-fields between the transmission line and the plate would tend to “cancel” each other and coupling efficiency would be degraded. Thus, any geometry that enhances the coupler structure performance based on magnetic field coupling must provide complementary magnetic fields with the same polarity as that distributed on the transmission line.
It is also apparent from FIG. 1 that E-fields are perpendicular to the H-field. Thus, for the E-fields to couple efficiently, the coupling structure should “overlap” the transmission line to maximize the conductor surface area which is perpendicular to the E-fields. This method of coupling can be described as Capacitive coupling (or broad edge coupling). This is different from a structure that utilizes magnetic field coupling, since maximizing perpendicular surface area to the H-field only requires a parallel structure to the transmission line, it does not necessitate overlap. These principles hold true for any three-dimensional (3-D) structure, including those which may be embedded into a PC board.
FIG. 2 illustrates a 3-D multi-plane coupling structure of the prior art with the associated upper and lower coupling plates 200 and 202 with current vectors (“io” and “i2”, respectively). The coupling plates 200 and 202 have an H-field generated by RF current “ii” propagating along transmission line 201 that induces H-fields of like polarity on coupling plates 200 and 202 for the surface facing the transmission. In other words, the H-field for the lower side of upper plate 200 is vectored the same as the top side of transmission line 201, and the H field for the top side of lower plate 202 is vectored the same as the lower surface of transmission line 201. These time varying magnetic fields generate current vectors io and i2 respectively. It is important to note that the H-field polarity of the lower surface of upper plate 200 CONFLICTS with the top surface polarity of lower plate 202. Thus, regions where the upper and lower plate overlap will cause reduced coupling efficiency, due to H-field cancellation, if they are electrically connected together (i.e. Current flows in the same direction in both structures). It is therefore imperative that the multi-layered helical geometry minimize regions of overlap between differing planes having current vectors oriented in the same direction.
From the foregoing discussion, it is evident that the two fields (E-field and H-field) are complementary, but independent mechanisms that can be used to provide coupling of RF energy. The electromagnetic relationship between them is determined by Maxwell's equations, which form guidelines that must be met to maximize coupling efficiency while providing superior directivity.
Directivity is a measure of the reflected and incident field differentiation of a coupler. The algebraic difference in decibels (dB) of the forward and reverse coupling coefficients for any fixed structure is defined as the directivity of that structure. A 20 dB directivity factor is considered acceptable for a bi-directional coupler. Historically, high performance bi-directional couplers have been fabricated on a substrate such as alumina, with thin film processes and their tight tolerance capability defining the coupler geometries to achieve a controlled 20 dB coupling coefficient with greater than 20 dB of directivity. Although the modularized approach to implementing the coupler is effective, it adds cost and process steps that could otherwise be eliminated if the coupler were to be embedded into the printed circuit board (PCB). To achieve high reliability coupler performance with existing PCB make tolerances of ±2 mil gaussian width distributions requires an innovative design approach. This innovation is based on selecting the proper coupling mode (E-field or H-field) that provides a design meeting manufacturing and performance requirements, while minimizing cost and area. Therefore, the need exists for embedded PCB coupler structure which achieves these desired objectives.